Fast bandwidth spectrum analysis

ABSTRACT

An apparatus includes a processor, a Phase-Locked Loop Waveform Generator (PLLWG), a Voltage Controlled Oscillator (VCO), a demodulator, signal conditioning circuitry, and an Analog-to-Digital Converter (ADC). The processor generates control command signals, receives a digital data input signal, and performs spectrum analysis on the digital data input signal. The PLLWG is coupled to the processor, receives the control command signals, and generates a charge pump output signal based on the control command signals. The VCO is coupled to the PLLWG, receives a tuning signal based on the charge pump output signal, and outputs a VCO output signal based on the tuning signal. The demodulator receives an incoming modulated signal and the VCO output signal, and outputs an analog output signal based on the incoming modulated signal and the VCO output signal. The ADC converts the analog output signal into the digital data input signal.

CROSS-REFERENCE TO RELATED APPLICATION

The present application is a continuation of U.S. patent applicationSer. No. 16/746,939 filed on Jan. 19, 2020, entitled “FAST BANDWIDTHSPECTRUM ANALYSIS”, the entire specification of which is herebyincorporated by reference in its entirety.

BACKGROUND OF THE DISCLOSURE 1. Field of the Disclosure

The disclosure relates in general to spectrum analysis, and moreparticularly, to fast narrow bandwidth spectrum analysis.

2. Background Art

Spectrum analysis entails measuring and reporting the magnitude of aninput signal as a function of its frequency. An example of such analysisincludes measuring the power level of a spectrum of known and unknownsignals. Spectrum analysis can be used to analyze electromagneticsignals over a defined band of frequencies. It can be used to determinethe frequencies of transmission sources, such as wireless networkingequipment, e.g., Wi-Fi and wireless routers, cellular equipment, dronesand their associated remote controllers, and other consumer, industrial,and military RF emitters.

SUMMARY OF THE DISCLOSURE

The disclosure is directed to an apparatus that is comprised of aprocessor, a Phase-Locked Loop Waveform Generator (PLLWG), a VoltageControlled Oscillator (VCO), a demodulator, and an Analog-to-DigitalConverter (ADC). The processor generates control command signals,receives a digital data input signal, and performs spectrum analysis onthe digital data input signal. The PLLWG is coupled to the processor,receives the control command signals, and generates a charge pump outputsignal based on the control command signals. The VCO is coupled to thePLLWG, receives a tuning signal based on the charge pump output signal,and outputs a VCO output signal based on the tuning signal. Thedemodulator receives an incoming modulated signal and the VCO outputsignal, and outputs an analog output signal based on the incomingmodulated signal and the VCO output signal. The ADC converts the analogoutput signal into the digital data input signal.

In some configurations, the VCO is a first VCO and the VCO output signalis a first VCO output signal, where the apparatus further comprises asecond VCO and a tuning signal. The second VCO is coupled to the PLLWGand receives the tuning signal based on the charge pump output signaland outputs a second VCO output signal based on the tuning signal. Thetuning signal switch receives the charge pump output signal andselectively outputs a first raw tuning signal and a second raw tuningsignal directed to the first VCO and the second VCO, respectively. Thedemodulator is an IQ demodulator that receives a selected frequencysignal based on one of the first and second VCO output signals, andoutputs an in-phase analog signal and a quadrature analog signal basedon the incoming modulated signal and one of the first and second VCOoutput signals. The ADC converts the in-phase analog signal and thequadrature analog signal into a digital data input signal, and outputsthe digital data input signal to the processor.

In some configurations, the tuning signal switch is a single-pole,double throw switch with an on-state resistance below 5 Ohms and anoff-state capacitance below 50 pF.

In some configurations, the apparatus further comprises a firstPhase-Locked Loop (PLL) filter and a second PLL filter. The first PLLfilter is coupled to both the PLLWG and the first VCO, and receives afirst raw tuning signal and conditions the first raw tuning signal. Thesecond PLL filter is coupled to both the PLLWG and the first VCO, andreceives a second raw tuning signal and conditions the second raw tuningsignal.

In some configurations, the apparatus further comprises an IQconditioner that receives an in-phase analog signal and a quadratureanalog signal from the IQ demodulator, and generates an in-phase analogdata signal and a quadrature analog data signal from the in-phase analogsignal and the quadrature analog signal, respectively.

In some configurations, the apparatus further comprises an imagerejection circuit that receives the in-phase analog data and thequadrature analog data from the IQ conditioner, provides a relativephase shift between the in-phase analog data and the quadrature analogdata to match the in-phase analog data and the quadrature analog data inphase while preserving relative amplitudes thereof, and sums thein-phase analog data and the quadrature analog data together to generatea combined analog data signal. Furthermore, the relative phase shiftbetween the in-phase analog data and the quadrature analog data ensuresthat, when summed, the image portion of the combined analog data signalis removed.

In some configurations, the apparatus further comprises a VCO selectswitch, coupled to the first VCO and the second VCO, to receive both thefirst VCO output signal and the second VCO output signal and selectivelyoutput one of the first VCO output signal and the second VCO outputsignal.

In some configurations, the VCO includes an oscillator element and a VCOamplifier. The oscillator element generates a radio frequency signal ofa frequency determined by a voltage level of the charge pump outputsignal and the VCO amplifier amplifies the radio frequency signal andoutputs the VCO output signal.

In some configurations, the VCO is one of a Maxim MAX2623 VCO and aMaxim MAX2622 VCO.

In some configurations, the processor is one of an NXP LPC43S70 with 16double-buffered 32-bit first-in-first-out Serial General PurposeInput/Output (SGPIO) pins capable of bit-shifting operation up to 102Mbps, a Field Programmable Gate Array (FPGA), and a System On a Chip(SOC).

The disclosure is also directed to a method that comprises generatingcontrol command signals, receiving a digital data input signal, andperforming spectrum analysis on the digital data input signal, by aprocessor, and receiving the control command signals and generating acharge pump output signal based on the control command signals, by aPLLWG coupled to the processor. The method further comprises receiving atuning signal based on the charge pump output signal and outputting aVCO output signal based on the tuning signal, by a VCO coupled to thePLLWG, and receiving an incoming modulated signal and the VCO outputsignal, and outputting an analog output signal based on the incomingmodulated signal and the VCO output signal, by a demodulator. The methodyet further comprises converting the analog output signal into thedigital data input signal, by an ADC.

In some configurations, the VCO is a first VCO, the VCO output signal isa first VCO output signal, and the demodulator is an IQ demodulator, themethod further comprising receiving the tuning signal based on thecharge pump output signal and outputting a second VCO output signalbased on the tuning signal, by a second VCO coupled to the PLLWG, andreceiving the charge pump output signal and selectively outputting afirst raw tuning signal and a second raw tuning signal directed to thefirst VCO and the second VCO, respectively, by a tuning signal switch.The method yet further comprises receiving a selected frequency signalbased on one of the first and second VCO output signals, and outputtingan in-phase analog signal and a quadrature analog signal based on theincoming modulated signal and one of the first and second VCO outputsignals, by the IQ demodulator, and converting the in-phase analogsignal and the quadrature analog signal into a digital data inputsignal, and outputting the digital data input signal to the processor,by the ADC.

In some configurations, the tuning signal switch of the method is asingle-pole, double throw switch with an on-state resistance below 5Ohms and an off-state capacitance below 50 pF.

In some configurations, the method further comprises receiving a firstraw tuning signal and conditioning the first raw tuning signal, by afirst Phase-Locked Loop (PLL) filter coupled to both the PLLWG and thefirst VCO, and receiving a second raw tuning signal and conditioning thesecond raw tuning signal, by a second PLL filter coupled to both thePLLWG and the first VCO.

In some configurations, the method further comprises receiving anin-phase analog signal and a quadrature analog signal from the IQdemodulator, and generating an in-phase analog data signal and aquadrature analog data signal from the in-phase analog signal and thequadrature analog signal, respectively, by an IQ conditioner.

In some configurations, the method further comprises receiving thein-phase analog data and the quadrature analog data from the IQconditioner, providing a relative phase shift between the in-phaseanalog data and the quadrature analog data to match the in-phase analogdata and the quadrature analog data in phase while preserving relativeamplitudes thereof, and summing the in-phase analog data and thequadrature analog data together to generate a combined analog datasignal, by an image rejection circuit. Furthermore, the relative phaseshift between the in-phase analog data and the quadrature analog dataensures that, when summed, the image portion of the combined analog datasignal is removed.

In some configurations, the method further comprises receiving both thefirst VCO output signal and the second VCO output signal and selectivelyoutput one of the first VCO output signal and the second VCO outputsignal, by a VCO select switch coupled to the first VCO and the secondVCO.

In some configurations, the method further comprises generating a radiofrequency signal of a frequency determined by a voltage level of thecharge pump output signal, by an oscillator element of the VCO, andamplifying the radio frequency signal and outputting the VCO outputsignal, by an amplifier of the VCO.

In some configurations, the VCO of the method is one of a Maxim MAX2623VCO and a Maxim MAX2622 VCO.

In some configurations, the processor of the method is one of an NXPLPC43S70 with 16 double-buffered 32-bit first-in-first-out SerialGeneral Purpose Input/Output (SGPIO) pins capable of bit-shiftingoperation up to 102 Mbps, a Field Programmable Gate Array (FPGA), and aSystem On a Chip (SOC).

BRIEF DESCRIPTION OF THE DRAWINGS

The disclosure will now be described with reference to the drawingswherein:

FIG. 1 illustrates an example apparatus to spectrum analyze an incomingmodulated signal, in accordance with at least one embodiment disclosedherein;

FIG. 2 illustrate an example graph showing frequencies stepped throughby the two VCOs of the apparatus of FIG. 1, in accordance with at leastone embodiment disclosed herein; and

FIG. 3 illustrates an example method for performing spectrum analysis onthe incoming modulated signal shown in FIG. 1, in accordance with atleast one embodiment disclosed herein.

DETAILED DESCRIPTION OF THE DISCLOSURE

While this disclosure is susceptible of embodiment(s) in many differentforms, there is shown in the drawings and described herein in detail aspecific embodiment(s) with the understanding that the presentdisclosure is to be considered as an exemplification and is not intendedto be limited to the embodiment(s) illustrated.

It will be understood that like or analogous elements and/or components,referred to herein, may be identified throughout the drawings by likereference characters. In addition, it will be understood that thedrawings are merely schematic representations of the invention, and someof the components may have been distorted from actual scale for purposesof pictorial clarity.

Referring now to the drawings and in particular to FIG. 1, an embodimentis disclosed that includes an apparatus 50 illustrated as including aplurality of components. The apparatus 50 receives a modulated RF signaland accurately measures a center frequency and occupied bandwidth of themodulated RF signal. Such capability is a potentially valuable aspect ofsignal analytics, whether or not the data content of the receivedmodulated RF signal itself is explicitly extracted. In at least oneembodiment, the apparatus 50 includes a pair of VCOs that are controlledby a single PLLWG, as discussed in more detail below. The apparatus 50permits a fast ramp to be performed over one VCO, then switched to asecond VCO with its own completely different fast ramp, all using a samemicroprocessor, PLLWG, and modulation conditioning components. This hasan advantage in a receiver subsystem that is expected to operate inshort windows of time-division-duplexed allocation. In the example ofFIG. 1, the apparatus 50 can perform receive operations in windows oftime that are, e.g., 2.5 μsec long, with one sampling event performedevery 25 μsec during transmit operations. This results in a 90% transmitduty cycle and 10% receive duty cycle that advances over two orders ofmagnitude faster than is possible with typical radio communicationsystems. The apparatus 50 has such advantages while being a relativelysimple apparatus, small, and low-powered and having rapid steppingcapability.

The timing constraints of the example of FIG. 1 are that one VCO must belocked and in use serving as the local oscillator for down-conversionduring the 2.5 μsec receive period. The other VCO is idle (i.e., notlocked or tuning) or turned off (e.g., to minimize coupled noise,harmonics, and/or spurious emissions into the receive chain) during thistime. Then when the apparatus 50 resumes transmit mode, the apparatus 50has ˜20 pec to either adjust the center frequency of the present VCO tothe next frequency value to receive a different subset of the same bandor to switch to the new VCO and lock frequency (at a minimum forreceived signal envelope detection) and/or phase (if also decodingdata). Typical PLL+VCO combinations cannot accomplish this timingrequirement and frequency flexibility with a tight size, cost, and powerbudget constraints.

The apparatus 50 includes a processor, such as microcontroller 100 thatgenerates control command signals 107 for the apparatus 50. In at leastone embodiment, the microcontroller 100 includes double-buffered pinswith bit shifting of at least 40 Mbps. In at least one embodiment, themicrocontroller 100 is an NXP LPC43S70 with 16 double-buffered 32-bitfirst-in-first-out SGPIO pins capable of bit-shifting operation up to102 Mbps. In other embodiment(s), one or more of any number of availablecomponents, such as a microprocessor(s), FPGA(s), or SOC(s) can performrole of the microcontroller 100.

The microcontroller 100 is coupled to a master clock generator 102 thatis also coupled to a PLLWG tuner 110 and an ADC 150. The master clockgenerator 102 generates a microcontroller clock signal 101 and outputsthis microcontroller clock signal 101 to the microcontroller 100. Themaster clock signal 102 is comprised of a plurality of synchronous clockgenerators, such as a first synchronous clock generator 155 thatgenerates the microcontroller clock signal 101 that first synchronousclock generator 155 outputs to the microcontroller 100, a secondsynchronous clock generator 156 that generates a PLLWG clock signal 111that the second synchronous clock generator 156 outputs to the PLLWGtuner 110, as well as a third synchronous clock generator 157 thatgenerates a variable ADC clock signal 151 that the third synchronousclock generator 157 outputs to the ADC 150. It is contemplated thatother timing resource configurations may be deployed in otherimplementations of the presently described subject matter as recognizedby one skilled in the art.

The microcontroller 100 generates control commands using high-speed orGPIO digital signal pins having a 1.8V logic “high” and a rate of speedlimited by its characteristics, the speed of its clock input, and thetime constant of the signal line and components. The microcontroller 100generates a step trigger signal 105 that instructs the PLLWG tuner 110to generate a charge pump output signal 112. In at least one embodiment,the PLLWG tuner 110 is further coupled to a tuning signal switch 113.The tuning signal switch 113 receives the charge pump output signal 112and switches the charge pump output signal 112 to one of a plurality ofoutputs, such as two outputs in the particular embodiment shown,although more outputs are contemplated. The tuning signal switch 113 isfurther coupled to a first PLL filter 120 and a second PLL filter 170.The tuning signal switch 113 outputs either a first raw tuning signal114 or a second raw tuning signal 115, the first raw tuning signal 114or the second raw tuning signal 115 being directed to a high-frequencyVCO 130 and a low-frequency VCO 180, respectively. The tuning signalswitch 113 outputs the first raw tuning signal 114 to the first PLLfilter 120, which conditions the first raw tuning signal 114 into afirst conditioned signal 125. Similarly, the tuning signal switch 113outputs the second raw tuning signal 115 to the second PLL filter 170,which conditions the second raw tuning signal 115 into a secondconditioned signal 175. In at least one embodiment, the tuning signalswitch 113 is a single-pole, double throw switch with an on-stateresistance below 5 Ohms and an off-state capacitance below 50 pF. In atleast one embodiment, the tuning signal switch 113 is an ONSemiconductor NLAS5123 single-pole, double throw switch with a 1.0 Ohmon-state resistance and a 20 pF off-state capacitance for an unusedoutput port when one output port is being used.

One skill in the art would recognize that a wide variety of switchingdevices and circuit configurations may be used to satisfy the functionalrequirements of the apparatus 50 to control the output of the PLLWG 110into or away from at least one VCO, such as a first VCO, e.g., thehigh-frequency VCO 130, and the second VCO, e.g., the low-frequency VCO180 shown, without departing from the scope of the embodimentsdisclosed. Depending upon the signal being detected by the apparatus 50,one skilled in the art would recognize that at least one embodiment ofthe apparatus 50 can utilize a single VCO, such as either thehigh-frequency VCO 130 or the low-frequency VCO 180, with associatedcomponents from a second VCO being likewise optionally omitted. Thoseskilled in the art of electronics design would recognize thatalternative switching devices and circuit configurations can be used inother implementations of the presently discussed subject matter, withoutdeparting from the scope of the embodiments disclosed. Likewise, thoseskilled in the art would recognize that a wide variety of on-state andoff-state resistive and reactive electrical characteristics will resultin these other implementations, as well as a wide variety of possibleand implemented timing characteristics to meet the requirements of eachapplication, without departing from the scope of the embodimentsdisclosed.

For each of the first and second raw tuning signals 114/115, either orboth if not selected to pass on to its PLL filter 120/170, respectively,have an Ohmic disconnect from the charge pump output signal 112 with alow parasitic series capacitance, with the dominant electricalcharacteristic being a high resistance path to analog reference ground.The purpose of this disconnect is to ensure a minimum of signal leakageto the unused path, and the purpose of the pre-set voltage is to providea deterministic voltage level as an input to the first PLL filter 120and to the second PLL filter 170 in such cases when they are notselected. It is recognized that these filtering circuits have apropensity for instability under certain operating conditions when theyhave a floating voltage for an input. In at least one embodiment, thetuning signal switch 113 has an off-state capacitance of only 20 pF,meaning only high frequency content of the charge pump output signal 112will pass through to the unselected port.

One skilled in the art would recognize that a wide variety of PLL filterdesigns may be used in other embodiments of the presently disclosedsubject matter, without departing from the scope of the embodiments.Some of these designs can include active architectures and some of whichcan include passive architectures. Each circuit can be comprised ofresistive and reactive components of various component values andorganization. In at least one embodiment in which only passivecomponents are utilized, a pre-set voltage can be omitted, as without anactive element, with the apparatus still being stable. In at least oneembodiment utilizing active components, that a variety of pre-setvoltages can be used, and others not requiring pre-set voltages at allfor embodiments that are unconditionally stable. Such electricalengineering details and opportunities for component selection,configuration, and reduction are left for those skilled in the art,without departing from the scope of the embodiments disclosed.

The first and second PLL filters 120/170 are coupled to first and secondVCOs, the high-frequency and low-frequency VCOs 130/180, respectively.The first conditioned signal 125 signal is used for tuning the outputfrequency of the first high-frequency VCO 130. In an analogous fashion,the second conditioned signal 175 is used for tuning the outputfrequency of the low-frequency VCO 180. The high-frequency VCO 130 ispowered by a first VCO voltage 131, and the low-frequency VCO 180 ispowered by a second VCO voltage 181, e.g., both 4.50 V. In at least oneembodiment, the high-frequency VCO 130 is a Maxim MAX2623 VCO, amonolithic microwave integrated circuit that tunes from 885 to 950 MHzbased on the input tuning voltage range of 0.4 to 2.4V and with outputpower that ranges between −14 and +0 dBm having phase noise of −101dBc/Hz at 100 kHz offset. In at least one embodiment, the low-frequencyVCO 180 is a Maxim MAX2622 VCO, which tunes from 855 to 881 MHz but hasotherwise similar characteristics. The output frequency can change at amoderate slew rate (˜100 MHz per microsecond) but only when presentedwith a strongly driven tuning signal. This reduced slew rate compared toother VCOs is the primary reason a buffering/integrating amplifier isincorporated into the PLL filters 120/170 to maintain stability at ahigh rate of tuning speed. One skilled in the art would recognize thatthe channel frequency, power level, voltage requirements, phase noise,slew rate, and other physical and electrical characteristics of otherembodiments of the presently described subject matter may besignificantly higher or lower based on the VCO used, without departingfrom the scope of the embodiments disclosed.

The high-frequency VCO 130 includes a first VCO oscillator 132 and afirst amplifier 134. The first VCO oscillator 132 receives the firstconditioned signal 125 and generates a first RF frequency signal 133that the first VCO oscillator 132 outputs to the first amplifier 134.The first amplifier 134 amplifies the first RF frequency signal 133 andgenerates a first amplified VCO output signal 135 which thehigh-frequency VCO 130 outputs. Similarly, the low-frequency VCO 180includes a second VCO oscillator 182 and a second amplifier 184. Thesecond VCO oscillator 182 received the second conditioned signal 175 andgenerates a second RF frequency signal 183 that the second VCOoscillator 182 outputs to the second amplifier 184. The second amplifier184 amplifies the second RF frequency signal 183 and generates a secondamplified VCO output signal 185 that the low-frequency VCO 180 outputs.In at least one embodiment, the low-frequency VCO is further coupled toa frequency divider 186. It is at this point where divergence occursbetween the high-frequency VCO 130 and low-frequency VCO 180, as thesecond amplifier 184 outputs the second amplified VCO output signal 185to the frequency divider 186. The frequency divider 186 receives thesecond amplified VCO output signal 185 and outputs a divided secondoutput signal 187.

In at least one embodiment, the frequency divider 186 is an AnalogDevices HMC432 monolithic microwave integrated circuit that divides itsfrequency input by two to generate an output having half the frequencyand a power level of −9 to −3 dBm, but otherwise similar electricalcharacteristics of its frequency input. In other embodiments of thepresently disclosed subject matter, other VCOs and/or frequency dividerscan be used. Similarly, frequency multipliers can be used, or nofrequency adjustment devices can be used. One skilled in the art wouldalso recognize that power splitters may be implemented immediately aftereach output signal is generated for feedback to the PLLWG, or before orafter a frequency adjustment or selection device in the chain withoutdivergence from the presently described subject matter. These minorvariations, as well as other amplification, attenuation, and/or matchingcircuit variations, are well-known and available to those skilled in theart of RF electronics design.

The high-frequency VCO 130 and the frequency divider 186 are bothfurther coupled to a VCO select switch 147. The VCO select switch 147receives both the first amplified VCO output signal 135 and a form ofthe second amplified VCO output signal 185, the divided second outputsignal 187, and selectively outputs one of the first amplified VCOoutput signal 135 and the divided second output signal 187. The VCOselect switch 147 is an RF switching component that presents zero or oneinput to its output, such as a selected frequency signal 145. The VCOselect switch 147 switches to a termination which is an input notselected for transmission, in at least one embodiment being a 50 Ohmbroadband resistive load to ground. In at least one embodiment, thistermination is designed into the VCO select switch 147 component itself.In at least one embodiment, the RF switch 147 is a PeregrineSemiconductor PE42420, a single-pole double throw switch with integrated50 Ohm termination to ground and a switching time of 300 nsec. In otherembodiments, many other types of switching components and/orterminations can be used without departing from the scope of thedisclosure.

The VCO select switch 147 is further coupled to an RF splitter 140. TheRF splitter 140 receives the selected frequency signal 145 andtransforms the selected frequency signal 145 into two signals, such as aselected frequency output signal 148 and a selected frequency feedback146. In at least one embodiment, the RF splitter 140 includes aProgrammable Logic Device (PLD) circuit. The RF splitter 140 outputs theselected frequency feedback 146 to the PLLWG tuner 110 that receives andmeasures, via the PLD circuit inside, the selected frequency feedback146 and matches it to a target output frequency. In at least oneembodiment, the apparatus 50 locks frequency in a time period of between0.5 and 3 pec for steps of 9.5 MHz. It is recognized that this timeperiod applies for operation of the example of FIG. 1 at any fastersampling rate up to and including 60 MSPS.

FIG. 2 illustrates a graph 200 showing frequencies stepped through bythe two VCOs of the apparatus 50 of FIG. 1. The graph 200 has a y-axis201 showing the amplitude of receive sensitivity and potential amplitudeof incoming signals. The x-axis 202 of the graph 200 shows the frequencyof the VCO center channels and incoming signals. The low-frequencybandwidth 203 covers a range of frequencies from 420 to 450 MHz,representing the U.S. ham radio band. The high-frequency bandwidth 233covers a range of frequencies from 902 to 928 MHz overlapping with theU.S. 900 MHz industrial, scientific, and Industrial, Scientific, andMedical (ISM) band. The low sub-bandwidth 204 illustrates the bandwidthof interest for each analytical step, representing 8 MHz of frequencycontent that will be sampled at 20 Msps. Note that the 20 Msps samplingcontains data up to 10 MHz subject to the Nyquist sampling criteria asknown to those skilled in the art of signal analytics. Similarly, thehigh sub-bandwidth 234 represents 9.5 MHz of frequency content that willbe sampled at 20 Msps.

The stepping pattern developed for the example of FIG. 1 is illustratedby the dashed arrows of FIG. 2. For the low-frequency bandwidth 203, thefirst low-frequency step 211 is at 418.00 MHz, which will provide afirst low-frequency input range 221, which includes frequency contentfrom 418 to 428 MHz. The first low-frequency input range 221 is shownwith filtering characteristics suggesting the very bottom and very topfrequencies taper off in amplitude. The result is that the bottom 2.00MHz and the top 2.00 MHz of frequency content will be discarded, and theanalysis will be performed on the content from 420.00 to 426.00 MHz. Ina similar manner, the second low-frequency step 212 at 427.00 MHz allowsthe apparatus 50 to receive a second low-frequency input range 222, thethird low-frequency step 213 at 435.00 MHz allows the apparatus 50 toreceive a third low-frequency input range 223, and the fourthlow-frequency step 214 at 443.00 MHz allows the apparatus 50 to receivea fourth low-frequency input range 224.

For the high-frequency bandwidth 233, the first high-frequency step 241is at 901.00 MHz, which will provide a first high-frequency input range251, which includes frequency content from 901 to 913 MHz. The sameconcept of the bottom 1.00 MHz and the top 2.50 MHz of frequency contentwill be discarded, and the analysis will be performed on the contentfrom 902.00 to 910.50 MHz. In a similar manner, the secondhigh-frequency step 242 at 910.50 MHz allows the apparatus 50 to receivea second high-frequency input range 252, the third high-frequency step243 at 920.00 MHz allows the apparatus 50 to receive a thirdhigh-frequency input range 223.

Several different step programs can be accessed by the microcontroller100 during operation to coincide with a demand for either fine or coarsesampling of received emissions in specific sub-bands. The master clocksignal 102 is variable and can accommodate changes in the ADC clocksignal 151 needed to change sampling speeds for wider or narrowerbandwidth as known to those skilled in the art of signal processing. Inat least one other embodiment of the presently described subject matter,other frequencies of preferred steps, bandwidths of interest, and/orpartial bands for rejection above, below, or in the interim bandwidthcan be used by those skilled in the art of signal processing to meet awide range of potential applications.

When the microcontroller 100 first triggers, by the step trigger signal105, the PLLWG tuner 110 to begin output suitable for coverage of thehigh-frequency bandwidth 233, the first frequency target is the firsthigh-frequency step 241, permitting the reception of the firsthigh-frequency input range 251. At the next triggering of step triggersignal 105, the PLLWG tuner 110 steps to the second high-frequency step242, permitting the reception of the second high-frequency input range252. This process repeats through a fourth high-frequency step (notshown), accomplished in a similar manner, whereupon additionally thePLLWG tuner 110 transmits a ramp complete signal 116 to themicrocontroller 100. The process for the coverage of the low-frequencybandwidth 203 is analogous, with the PLLWG tuner 110 generating sameramp complete signal 116 to the microcontroller 100 when complete. Notethat in at least one embodiment utilizing two PLLWGs, the channel centerfrequencies detailed in each ramp's definition in the first PLLWG andsecond PLLWG can be completely different, with different start, stop,step size, and/or number of steps. These ramp definitions can alsochange in one or both of the PLLWGs during operation as required.

In at least one embodiment, the example of FIG. 1 and FIG. 2 can usefrequency selection data as part of the step trigger signal 105 todefine a center frequency in a conventional manner than is provided forin the stepped ramp approach. The time it takes to lock onto a newfrequency in this mode is much longer than with other modes. However,this provides for a means to “zoom in” to a signal identified in acoarse scan by selecting a frequency immediately below the frequency ofinterest without having to cycle through steps of a many-step ramp toget to a single frequency location of interest.

In at least one other embodiment of the presently disclosed subjectmatter, the main pre-selected channels will not be progressed in amonotonic upwards or downwards fashion, and instead will vary based onthe presence of target radio activity that must be monitored along withscanning through the rest of the band to detect new activity in otherparts of each band. Similarly, the manner in which demodulation occurswill also vary in certain embodiment(s) of the presently disclosedsubject matter, so that demodulation with coarse signals having few datapoints may be interspersed with fine signal analytics having many moredata points. This permits finer analysis of precise channel occupancy bytarget radio emissions, as well as modulation type and subcarrierspacing. These and other transmission characteristics may be of interestin a variety of applications where rapidly adaptive scanning anddemodulation is of intrinsic value.

Once the selected frequency output signal 148 of the example apparatus50 of FIG. 1 has been selected, the operation continues. The RF splitter140 is further coupled to an IQ demodulator 160 shown as also beingcoupled to an IQ conditioner 163. The IQ demodulator 160 receives anincoming modulated signal 165 (the signal to be analyzed by theapparatus 50) as one input, with the selected frequency signal 148,known as the local oscillator for the demodulation circuit, beingreceived as a second input from the RF splitter 140. The IQ conditioner163 receives an in-phase analog output signal 161 and a quadratureanalog output signal 162 from the IQ demodulator 160, two signals thatcontain the data that had previously been incorporated into the incomingRF signal of interest. The IQ conditioner 163 generates an in-phaseanalog data signal 164 and a quadrature analog data signal 152 from thein-phase analog output signal 161 and the quadrature analog outputsignal 162, respectively. Such conditioning in the analog domaininvolves a switchable amplification stage, multi-pole filtering, andtunable amplification as a driver stage prior to introduction to animage reject circuit. In at least one embodiment, the IQ conditioner 163is an Analog Devices ADRF6510, a monolithically integrated circuitcombining a matched pair of differential switched gain amplifiers,programmable low-distortion 6-pole filters for anti-aliasing, andlow-noise amplification/attenuation stages for each of the in-phase andquadrature data channels, suitable for directly driving operationalamplifiers of an image reject circuit or an ADC. In at least one otherembodiment, a wide variety of IQ conditioning circuitry can be used.Examples of potential IQ conditioning circuitry known to those skilledin the art of signal analytics include passive filters, active filters,switched gain stages, low noise amplifiers, variable attenuators usingswitched resistive networks, variable amplifiers using switched gainnetworks, and other fixed and tunable signal conditioning circuitsavailable to those skilled in the art of RF and analog electronicsdesign. In at least one embodiment, a low pass filter (not shown), suchas a programmable anti-alias filter, is disposed before the ADC 150. Inat least one embodiment, the IQ conditioner 163 includes such a low passfilter. In at least one embodiment, the IQ conditioner 163 includes asignal gain control.

In at least one embodiment, the IQ conditioner 163 is further coupled toan image reject circuit 166. The image reject circuit 166 receives thein-phase analog data signal 164 and the quadrature analog data signal152 from the IQ conditioner 163. The image reject circuit 166 provides arelative phase shift between the two data streams using operationalamplifier circuitry to match the real signal content in phase whilepreserving relative amplitudes, matching the image signal content 180degrees out of phase while preserving relative amplitudes, then sums thetwo signals together into a single output as a combined analog datasignal 167. The relative phase shift between the in-phase analog dataand the quadrature analog data ensures that, when summed, the imageportion of the combined analog data signal is removed. This provides asingle (stronger) real data signal containing the amplitude data of theoriginal signal at a single sideband of real content while eliminatingthe second sideband of image content. Single-sideband conversion is asignificant advantage well known to those skilled in the art of signalanalytics. In at least one other embodiment of the presently discussedsubject matter, instead of combining the in-phase and quadrature data inthe analog domain these signals are phase shifted and then compensatedfor in the digital domain to eliminate the image content. This has theadvantage of eliminating the analog image reject circuit for a savingsin circuit board space and layout complexity, with the disadvantage ofrequiring additional digital signal processing and power consumption bycomparison. In at least one embodiment, additional ADC and signalprocessing resources may not be substantially disadvantageous, whereasin other implementations, such as in the example of FIG. 1, suchresource requirements were neither available nor desirable toincorporate. In the example of FIG. 1, the image reject circuit 166 iscomprised of a three-stage matched-pair operational amplifier circuit, asumming circuit, a differential-to-single ended signal conversioncircuit, and a level shifting circuit to present a final output with anon-zero common mode suitable for driving into the ADC 150. In at leastone embodiment, the apparatus 50 can omit the image rejection circuit166, instead utilizing separate ADC converters (not shown) for the I andQ data. In such an embodiment(s), image rejection is performed by anFPGA (not shown) or by a processor, such as the microcontroller 100.

The image rejection circuit 166 is further coupled to an ADC 150. In atleast one embodiment, the ADC 150 receives the combined analog datasignal 167 that includes information from the in-phase analog outputsignal 161 and the quadrature analog output signal 162 while rejectingthe imaged content. The ADC 150 also receives the ADC clock signal 151to sample analog data, store the sample results in an internal databuffer, and generate a digital data input signal 106. The ADC 150outputs the digital data input signal 106, with the microcontroller 100receiving the digital data input signal 106 and performing spectrumanalysis on the digital data input signal 106. The microcontroller 100can perform any number of different types of spectrum analysis on thedigital data input signal 106, as known to those skilled in the art. Inat least one embodiment, the ADC 150 can be part of a SOC, or, in atleast one embodiment can be a discrete single or dual channel ADC. In atleast one embodiment, the ADC 150 can include a plurality of channelswith the ADC 150 utilizing a channel(s) as needed for a particularapplication.

A wide variety of ADCs can be used for the equivalent function in otherimplementations of the presently discussed subject matter, withoutdeparting from the scope of the disclosure. Such variety includes ADCsas separate components as in the present example of FIG. 1, but ofdifferent makes and models and performance characteristics. Such varietyis also contemplated to include ADCs integrated internally with themicroprocessors and other computing devices controlling eachimplementation or multiple implementations in a parallel fashion. Othercombinations of ADC functionality may be programmed in the firmware ofan FPGA, SOC, or other programmable computing device. Yet otherembodiments are contemplated to incorporate the ADC function into alarger integrated circuit that includes the down-converting mixer,filtering, amplification, and/or image reject functions as available inthe integrated component marketplace or in custom designs by thoseskilled in the art of RF and analog electronics design.

FIG. 3 illustrates a method 300 including a process 310 generating thecontrol command signals 107, receiving the digital data input signal106, and performing spectrum analysis on the digital data input signal106. In at least one embodiment, process 310 is performed by aprocessor, such as the microcontroller 100. Process 310 proceeds toprocess 320.

Process 320 includes receiving the control command signals 107 andgenerating the charge pump output signal 112 based on the controlcommand signals 107. In at least one embodiment, this process 320 isperformed by the PLLWG 110 coupled to the processor 100. Process 320proceeds to process 330.

Process 330 includes receiving a tuning signal, such as at least one ofthe first raw tuning signal 114 and the second raw tuning signal 115,based on the charge pump output signal 112 and outputting a VCO outputsignal, such as at least one of the amplified first VCO output signal135 and the second amplified VCO output signal 185. based on this tuningsignal. In at least one embodiment, this process 330 is performed by aVCO, such as by at least one of the high-frequency VCO 130 and thelow-frequency VCO 180 that are coupled to the PLLWG 110. Process 330proceeds to process 340.

Process 340 includes receiving the incoming modulated signal 165 and theVCO output signal from process 330, and outputting an analog outputsignal, such as at least one of the in-phase analog output signal 161and the quadrature analog output signal 162, based on the incomingmodulated signal 165 and the VCO output signa from process 330. In atleast one embodiment, this process 340 is performed by a demodulator,such as the IQ demodulator 160. Process 340 proceeds to process 350.

Process 350 includes converting the analog output signal from process340 into the digital data input signal 106. In at least one embodiment,process 350 is performed by the ADC 150. The method 300 can furtherinclude any of the processes and any of the components of the apparatus50 described above for FIG. 1.

The foregoing description merely explains and illustrates the disclosureand the disclosure is not limited thereto except insofar as the appendedclaims are so limited, as those skilled in the art who have thedisclosure before them will be able to make modifications withoutdeparting from the scope of the disclosure.

1-20. (canceled)
 21. An apparatus, comprising: a processor to generatecontrol command signals, receive a digital data input signal, andanalyze the digital data input signal; a Phase-Locked Loop WaveformGenerator (PLLWG), coupled to the processor, to receive the controlcommand signals and generate a charge pump output signal based on thecontrol command signals; a Voltage Controlled Oscillator (VCO), coupledto the PLLWG, to receive a tuning signal based on the charge pump outputsignal and output a VCO output signal based on the tuning signal; ademodulator to receive an incoming modulated signal and the VCO outputsignal, and output an in-phase analog data signal and a quadratureanalog data signal based on the incoming modulated signal and the VCOoutput signal; an image reject circuit to receive the in-phase analogdata signal and the quadrature analog data signal and sum the in-phaseanalog data signal and the quadrature analog data signal into the analogoutput signal; and an Analog-to-Digital Converter (ADC) to convert theanalog output signal into the digital data input signal.
 22. Theapparatus according to claim 21, wherein the VCO is a first VCO and theVCO output signal is a first VCO output signal, the apparatus furthercomprising: a second VCO, coupled to the PLLWG, to receive the tuningsignal based on the charge pump output signal and output a second VCOoutput signal based on the tuning signal; and a tuning signal switch toreceive the charge pump output signal and selectively output a first rawtuning signal and a second raw tuning signal directed to the first VCOand the second VCO, respectively; wherein the demodulator is an IQdemodulator to receive a selected frequency signal based on one of thefirst and second VCO output signals, and output the in-phase analog datasignal and the quadrature analog data signal based on the incomingmodulated signal and one of the first and second VCO output signals; andwherein the ADC converts the in-phase analog data signal and thequadrature analog data signal into a digital data input signal, andoutputs the digital data input signal to the processor.
 23. Theapparatus according to claim 22, wherein the tuning signal switch is asingle-pole, double throw switch with an on-state resistance below 5Ohms and an off-state capacitance below 50 pF.
 24. The apparatusaccording to claim 22, further comprising: a first Phase-Locked Loop(PLL) filter coupled to both the PLLWG and the first VCO, the first PLLfilter to receive a first raw tuning signal and condition the first rawtuning signal; and a second PLL filter coupled to both the PLLWG and thefirst VCO, the second PLL filter to receive a second raw tuning signaland condition the second raw tuning signal.
 25. The apparatus accordingto claim 22, further comprising an IQ conditioner to receive thein-phase analog signal and the quadrature analog signal from the IQdemodulator, and to generate an in-phase analog data signal and aquadrature analog data signal from the in-phase analog signal and thequadrature analog signal, respectively.
 26. The apparatus according toclaim 25, further comprising an image rejection circuit to receive thein-phase analog data and the quadrature analog data from the IQconditioner, provide a relative phase shift between the in-phase analogdata and the quadrature analog data to match the in-phase analog dataand the quadrature analog data in phase while preserving relativeamplitudes thereof, and sum the in-phase analog data and the quadratureanalog data together to generate a combined analog data signal, therelative phase shift between the in-phase analog data and the quadratureanalog data ensuring that, when summed, an image portion of the combinedanalog data signal is removed.
 27. The apparatus according to claim 22,further comprising a VCO select switch, coupled to the first VCO and thesecond VCO, to receive both the first VCO output signal and the secondVCO output signal and selectively output one of the first VCO outputsignal and the second VCO output signal.
 28. The apparatus according toclaim 21, wherein the VCO includes an oscillator element and a VCOamplifier, the oscillator element to generate a radio frequency signalof a frequency determined by a voltage level of the charge pump outputsignal and the VCO amplifier to amplify the radio frequency signal andoutput the VCO output signal.
 29. The apparatus according to claim 21,wherein the VCO is one of a Maxim MAX2623 VCO and a Maxim MAX2622 VCO.30. The apparatus according to claim 21, wherein the processor is an NXPLPC43S70 with 16 double-buffered 32-bit first-in-first-out SerialGeneral Purpose Input/Output (SGPIO) pins capable of bit-shiftingoperation up to 102 Mbps.
 31. A method, comprising: generating controlcommand signals, receiving a digital data input signal, and analyzingthe digital data input signal, by a processor; receiving the controlcommand signals and generating a charge pump output signal based on thecontrol command signals, by a Phase-Locked Loop Waveform Generator(PLLWG) coupled to the processor; receiving a tuning signal based on thecharge pump output signal and outputting a voltage controlled oscillator(VCO) output signal based on the tuning signal, by a VCO coupled to thePLLWG; receiving an incoming modulated signal and the VCO output signal,and outputting an in-phase analog data signal and a quadrature analogdata signal based on the incoming modulated signal and the VCO outputsignal, by a demodulator; receiving the in-phase analog data signal andthe quadrature analog data signal and summing the in-phase analog datasignal and the quadrature analog data signal into the analog outputsignal, by an image reject circuit; and converting the analog outputsignal into the digital data input signal, by an Analog-to-DigitalConverter (ADC).
 32. The method according to claim 31, wherein the VCOis a first VCO, the VCO output signal is a first VCO output signal, andthe demodulator is an IQ demodulator, the method further comprising:receiving the tuning signal based on the charge pump output signal andoutputting a second VCO output signal based on the tuning signal, by asecond VCO coupled to the PLLWG; receiving the charge pump output signaland selectively outputting a first raw tuning signal and a second rawtuning signal directed to the first VCO and the second VCO,respectively, by a tuning signal switch; receiving a selected frequencysignal based on one of the first and second VCO output signals, andoutputting the in-phase analog data signal and the quadrature analogdata signal based on the incoming modulated signal and one of the firstand second VCO output signals, by the IQ demodulator; and converting thein-phase analog signal and the quadrature analog signal into a digitaldata input signal, and outputting the digital data input signal to theprocessor, by the ADC.
 33. The method according to claim 32, wherein thetuning signal switch is a single-pole, double throw switch with anon-state resistance below 5 Ohms and an off-state capacitance below 50pF.
 34. The method according to claim 32, further comprising: receivinga first raw tuning signal and conditioning the first raw tuning signal,by a first Phase-Locked Loop (PLL) filter coupled to both the PLLWG andthe first VCO; and receiving a second raw tuning signal and conditioningthe second raw tuning signal, by a second PLL filter coupled to both thePLLWG and the first VCO.
 35. The method according to claim 32, furthercomprising receiving an in-phase analog signal and a quadrature analogsignal from the IQ demodulator, and generating the in-phase analog datasignal and the quadrature analog data signal from the in-phase analogsignal and the quadrature analog signal, respectively, by an IQconditioner.
 36. The method according to claim 35, further comprisingreceiving the in-phase analog data and the quadrature analog data fromthe IQ conditioner, providing a relative phase shift between thein-phase analog data and the quadrature analog data to match thein-phase analog data and the quadrature analog data in phase whilepreserving relative amplitudes thereof, and summing the in-phase analogdata and the quadrature analog data together to generate a combinedanalog data signal, by an image rejection circuit, the relative phaseshift between the in-phase analog data and the quadrature analog dataensuring that, when summed, an image portion of the combined analog datasignal is removed.
 37. The method according to claim 32, furthercomprising receiving both the first VCO output signal and the second VCOoutput signal and selectively output one of the first VCO output signaland the second VCO output signal, by a VCO select switch coupled to thefirst VCO and the second VCO.
 38. The method according to claim 31,further comprising: generating a radio frequency signal of a frequencydetermined by a voltage level of the charge pump output signal, by anoscillator element of the VCO; and amplifying the radio frequency signaland outputting the VCO output signal, by an amplifier of the VCO. 39.The method according to claim 31, wherein the VCO is one of a MaximMAX2623 VCO and a Maxim MAX2622 VCO.
 40. The method according to claim31, wherein the processor is an NXP LPC43S70 with 16 double-buffered32-bit first-in-first-out Serial General Purpose Input/Output (SGPIO)pins capable of bit-shifting operation up to 102 Mbps.